There have been several attempts at a triode driver for the QUAD II KT66 output stage ~ since 1980 I have tried 'a few' myself ~ In every case I encountered problems that had already been solved by the elegant QUAD EF86 circuit so there had to be a good reason to change to a triode input stage and why QUAD did not use triodes

Changes to a triode driver will often require at least 3 triodes or a pentode plus 2 triodes ~ If you use ECCnn the heater current will increase (but not double) an EF86 has 200mA heater current @ 6.3Va.c. and ECC81 or ECC83 300mA ~ ECC88 and some others require over 350mA per valve and will further load the mains transformer

Some changes like the Glen Croft 6080 HT regulator and the so called concordant modifications can overload the mains transformer even before adding the load of a QUAD 22 pre-amp and tuners despite the replacement of the GZ32 with silicon diode rectifiers to give a higher HT which causes higher internal heat dissipation

To get enough gain for the 50V to 60Va.c. push pull signal at the KT66 grid1's for a 15W output appears to require 2 series stages of triode amplification which would be less stable than a single stage EF86 driver when at least 15dB overall negative feedback has to be applied to meet the distortion spec of the QUAD KT66 output stage

The circuit above uses a long tail or cathode coupled pair with the tail resistor R12 the same value as the load resistors R5 and R6 which are 180kΩ ~ R7 and R9 are 680kΩ which are the same values as the QUAD II circuit so the only difference in the two circuits from V2 anodes will be the internal anode resistance r_a of each half of V2

Previously had I used only a long tail pair with various high mu triodes to try and simulate the single stage QUAD II driver design ~ So that negative feedback (NFB) from the output transformer can be directly applied to the long tailed pair non inverting input grid ~ both control grids (g1) ideally need to be referenced to 0V

HT3 is a negative supply derived from a pair of silicon diodes in full wave rectifier across the HT winding ~ The positive HT via V5 and L1 is about 40V to 60V lower than the negative HT3 ~ R14 is chosen to position the anodes of V2 for maximum unclipped swing with minimum distortion while also providing additional smoothing with C7

I had also tried semiconductor current sources in place of R12 without a –Ve supply and current source loads for R5 and R6 but always more open loop gain was required even if only a small amount of overall NFB was to be applied and always the circuit was too complicated for little if any performance advantage ~ Welcome to triode µ

With the values show above the anodes of V2 sit at about 170V and the gain from V2 input to each output was about 30x to 35x for various samples of ECC81 ~ When coupled to the KT66 g1's ≈1.2V to 1.4V is required for a 10W output ~ The overall gain from C1 input to speaker terminal output is about 6.5x and 1.7V is required for 15W output (11V across 8Ω or 15V across 15Ω) but this is before feedback is applied

When fitted with ECC83s the gain is higher at ≈15x but no bias adjustments are required and it is this feature of the long tail pair with –ve supply and direct NFB that I liked ~ The circuit can be used with almost any pin compatible valves but some like ECC82 will require >5Vinput for 15W out and with NFB the input may need to be greater still

The mutual conductance g_m of triode valves increases with anode current I_a but the value of load resistors R_L {R5 and R6} will need to be lowered to maintain the operating point ~ The internal anode resistance r_a decreases with I_a and because it appears in parallel with R_L the gain of a practical long tail pair amplifier may not be much higher

As I_a increases ~ the increase in g_m and the reduction in r_a tend to 'level out' ~ as does the maximum possible gain µ available from a triode ~ There appears to be little advantage using high I_a in a triode voltage amplifier because the power dissipated in the valves and the load resistors increases greater than the gain

Many triodes can become more linear as I_a increases and because ∆r_a and ∆g_m level out with ∆I_a the effect of dynamic changes also reduces The Triode Sound ~ To achieve maximum gain with high with I_a ~ current sources could be used for the loads but this adds yet more complexity to obtain a stable operating point

The long tail pair circuit was modelled with PSPICE and some manual calculations were to done to establish how it would interface with the QUAD KT66 output stage but the only way to tell for sure was to build the circuit and fit it to a QUAD II in place of the EF86 driver and measure the open loop gain and frequency response etc.

It is a single sided board with PCB mounted ceramic valve holders fitted on the solder side as used for my other valve project PCBs ~ I have seveal PCBs still available if you want to try the circuit

Capacitors C1 ~ C2 ~ C3 are mounted on turret tags to enable different types to be easily changed without having to remove the PCB

It would appear that the QUAD II was designed for about 40dB or 100x open loop gain and for 20dB or 10x closed loop gain ~ The feedback is taken from point Q on the OPT which is at half the output voltage for the 15Ω tap and the 'amount of feedback' as traditionally calculated by 20Log(1+ßA_o) is 20Log(1+0.175x50) = 19.8dB

Applying more feedback to reduce A_o by 27dB can show signs of instability and this is typical of many valve amplifiers ~ The gain margin needs to be at least 6dB to allow for open circuit stability and not much more as A_o will not be large to start with ~ Beware of any audio amplifier that 'Must not be operated without a load'

As marked on the schematic above the aim was to apply overall F.B. to the non inverting grid of V₂ but the voltage gain of the output stage from each KT66 g1 to the 15Ω output is about 0.5x or –6dB so the 25dB gain from the long tailed pair phase splitter using ECC81 gave only 20dB of open loop gain requiring 100% F.B. by making R11 0Ω to achieve similar output impedance and performance to the QUAD circuit

The long tailed pair phase splitter with negative supply will accept many different valves without bias changes and this was interesting because apart from slight gain variations there appears to be no difference in sound between ECC81s ECC82s and ECC83s so no good for a valve shootout ~ There is a sharper stereo image and possibly better control of transients so maybe triodes do have something that pentodes don't ~ In some configurations

The input gain stage V1 is based on my amplifier/buffer gain block using an un–bypassed cathode resistor and positive raised g1 for the first valve but to achieve 20dB gain to drive the cathode coupled phase splitter I used the 'auto bias' value for R4 with R2 not fitted and adjusted R3 for 20dB gain and R16 for the operating point

The reason the amplifier buffer configuration was chosen was it offers a predictable gain outside the phase splitter feedback loop which is simply determined by the ratio of R₃ and R₄ ~ The cathode follower ensures this gain is not affected by loading and its low output impedance ensures the Miller capacitance of the splitter is not a problem

The driver performed well but the valve for V₁ needs to be an ECC81 and different ECC81s showed that sometimes the third harmonic was higher than the second and although this was not apparent from listening it was something I wanted to avoid plus the primary source of distortion was from this first stage so back to . . .

How does the QUAD II EF86 Driver work so well

The additional large negative supply required for the cathode coupled phase splitter is not difficult to arrange but the original QUAD driver design was simpler and achieves about 46dB gain from input to each KT66 g1 with only 2 pentodes and a single HT well smoothed supply

I had previously used triode cascodes in a cathode coupled phase splitter circuit with a 350V negative supply similar to the one above and also in the QUAD driver circuit topology which ~ along with a html version of Amplifiers and Superlatives ~ was the first 'content' for my original 1993 dc~daylight website

If the circuit were built as a traditional cathode coupled phase splitter ~ with R8 short circuited ~ the gain from input to V3 g1 would be about 43dB and to V4 g1 via cathode coupling would be about 37dB ~ Without using a large negative supply and a large 'tail' resistor the way to adjust the drive balance would be to make R6 about 500kΩ

Clearly the imbalance between R5 and R6 would be too great to allow sensible d.c. bias conditions and is also impractical because the frequency response would be very different from each each EF86 ~ The balanced drive would be about 3.5dB less than could be achieved with a circuit like the simple paraphase splitter which could be used because the KT66 valves draw no g1 current and are auto biased

Note in the QUAD II circuit the d.c. path for the KT66 control grids through R7 and R9 is not directly connected to 0V ground but is via R10 in parallel with R11 and because the cathode currents of V1 and V2 pass through these resistors the KT66 grids will be at ≈200mV not 0V even with perfect coupling capacitors C2 and C3

R7 and R9 now feed g1 of V2 with a signal which is the 'difference' between the outputs of V1 and V2 and as such the circuit is self balancing as described here

The common cathode resistors R4+R10 carry the equal and opposite currents of V1 and V2 and these cancel causing zero voltage change at the connected cathodes so the maximum gain available from V1 is achieved as if the resistors were bypassed to ground

So why not use this circuit?

The main problem is however when applying overall negative feedback from the output transformer ~ Negative feedback needs to be applied to V1 and not V2 or the circuit will be unstable ~ The feedback could be applied to g1 of V1 but not easily without lowering the input imedpance and the loading the 22 pre–amp

If the feedback were applied to the junction R4 R10 without some circuit changes it would be negative feedback for V1 but positive for V2 ~ The change in the QUAD II design was to place a low value resistor R8 between R7 and R9 such that the signal across R8 was just enough to give V2 the same but opposite phase output as V1

The QUAD circuit is now similar to a regular paraphase design where the junction of R8 and R9 would be connected to signal ground and (R8+R7)/R8 would match the gain of V1 suggesting a gain of 48dB for each output ~ This allows for a lower resistance path for the KT66 g1's without undue loading on the splitter outputs

The junction of R8 and R9 in the QUAD II is connected to the junction of R4 R10 and the feedback resistor R11 ~ The feedback signal is developed across R10 and the input to V2 across R8+R4 ~ R8 is 'floating' between R7 and R9 so any signal at the junction of R4 and R10 also appears at V2 g1 so V2 is not subject to positive or negative feedback

The QUAD II driver is a simple cathode coupled paraphase phase splitter with additional a.c. coupling via C1 between the screen grids (g2) of the EF86 pentodes ~ The clever bit is preventing the positive feedback to V2 by positioning the junction of R8 and R9 at the feedback point and not to signal ground

With coupling between cathodes as well as the paraphase drive to V2 the C1 coupling between the screen grids of the EF86s may not be so desirable ~ The effect of this coupling could be eliminated by returning each EF86 g2 to its cathode with a larger value capacitor ~ A triode driver would not have this coupling so does C1 affect the sound ?

An obvious choice to directly replace the pentodes in the QUAD design or other is a pair of cascaded triodes acting as a single valve in the configuration known as a cascode

Cascodes using dual triodes like the ECC81 and ECC83 operated at I_a ≈1mA will give >46dB of gain using the same values for R5 R6 R7 and R9 as the QUAD circuit

The schematic on the right is a variation of my 1992 cascode driver with the addition of R14 to balance the d.c. operating point of V2 and this time using ECC88s in place of the ECC81s

Building the cascode driver shown above using the original tag board is quite simple and makes for a much neater final product than some other triode mods ~ Note the top anode and bottom cathode of the cascode are the same pins as the EF86 but the wire between pins 3 and 8 of V1 and V2 have to be removed to join cathodes

One problem with the cascode ~ especially when using an ECC88 ~ was despite using well balanced dual triodes there was often a large d.c. imbalance at the output anodes ~ This problem should also be seen on a standard QUAD II using EF86s because the auto bias voltage for V1 is across R4+(R10||R11) and the bias for V2 is only across R4

Whether using cascode triodes or EF86s the imbalance can be compensated for by inserting a resistor R14= 2x(R10||R11) in series with V2 cathode ~ R14 also tends to linearise the V2 cascode but also lowers its open loop gain ~ The open loop balance of the outputs can be restored by increasing R8 to about 3k3Ω to 4k7Ω

Some ECC88 6DJ8 and E88CC data sheets state that the grounded cathode stage ~ bottom valve ~ should be pins 6 7 and 8 or section a' g' and k' ~ Other experts on the web further complicate matters stating that the heater to cathode insulation is lower for h–k' which is not true it is the same for both sections and the use of pins 6 7 and 8 for the common cathode section relates to making an optimum VHF amplifier so can be ignored for audio

I have made a double sided PCB with ground–plane for the cascode mod which has been fitted in my Majestic Mod QUAD IIs ~ In all my triode 'mods' I have tried to keep the anode loads and supplies around the valves similar to those of the EF86 circuit such that only the features intrinsic to the triodes are being evaluated

With the E88CC cascode it is possible to use the original QUAD circuit topology with minimum changes ~ Lowering the anode load resistors to about 100kΩ and operating each valve at about 2mA gives a 'tighter' drive to the KT66 grids and lower overall phase shift to 100kHz and above ~ increasing the anode current beyond 2mA has little advantage because the load resistors R5 and R6 need to be further reduced to maintain the d.c. conditions